Processing for improved performance and reduced pilot

ABSTRACT

A digital spread spectrum communication system employing pilot-aided coherent multipath demodulation effects a substantial reduction in global-pilot and assigned-pilot overheads. The system and method uses a QPSK-modulated data signal whereby the modulated data is removed and the recovered carrier is used for channel amplitude and phase estimation. The resulting signal has no data modulation and is used as a pseudo-pilot signal. In conjunction with the pseudo-pilot signal, a multiple-input phase-locked loop is employed further eliminating errors due to carrier-offset by using a plurality of pseudo-pilot signals. A pilot signal is required to resolve absolute phase ambiguity, but at a greatly reduced magnitude.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of application Ser. No. 09/078,417,filed on May 14, 1998. Now U.S. Pat. No. 6,366,607.

BACKGROUND

1. Field of the Invention

The present invention relates generally to digital communications. Morespecifically, the invention relates to a system for and method of usinga code division multiple access air interface which greatly reduces thesignal power required for the global and assigned-pilots while improvingperformance by using the quadrature phase shift keyed (QPSK) trafficsignal for a particular channel to perform channel estimation andcarrier recovery.

2. Description of the Prior Art

Most advanced communication technology today makes use of digital spreadspectrum modulation or code divisional multiple access (CDMA). Digitalspread spectrum is a communication technique in which data istransmitted with a broadened band (spread spectrum) by modulating thedata to be transmitted with a pseudo-noise signal. CDMA can transmitdata without being affected by signal distortion or an interferingfrequency in the transmission path.

Shown in FIG. 1 is a simplified CDMA communication system that involvesa single communication channel of a given bandwidth which is mixed by aspreading code which repeats a predetermined pattern generated by apseudo-noise (pn) sequence generator. A data signal is modulated withthe pn sequence producing a digital spread spectrum signal. A carriersignal is then modulated with the digital spread spectrum signalestablishing a forward link, and transmitted. A receiver demodulates thetransmission extracting the digital spread spectrum signal. Thetransmitted data is reproduced after correlation with the matching pnsequence. The same process is repeated to establish a reverse link.

During terrestrial communication, a transmitted signal is disturbed byreflection due to varying terrain and environmental conditions andman-made obstructions. This produces a plurality of received signalswith differing time delays at the receiver. This effect is commonlyknown as multipath propagation. Moreover, each path arrives delayed atthe receiver with a unique amplitude and carrier phase.

To identify the multiple components in the multipath propagation, therelative delays and amplitudes and phases must be determined. Thisdetermination can be performed with a modulated data signal, buttypically, a more precise rendering is obtained when compared to anunmodulated signal. In most digital spread spectrum systems, it is moreeffective to use an unmodulated pilot signal discrete from thetransmitted modulated data by assigning the pilot an individual pnsequence. A global-pilot signal is most valuable on systems where manysignals are transmitted from a base station to multiple users.

In the case of a base station which is transmitting many channels, theglobal-pilot signal provides the same pilot sequence to the plurality ofusers serviced by that particular base station and is used for theinitial acquisition of an individual user and for the user to obtainchannel-estimates for coherent reception and for the combining of themultipath components. However, at the required signal strength, theglobal-pilot signal may use up to 10 percent of the forward directionair capacity.

Similar multipath distortion affects a user's reverse link transmissionto the base station. Inserting in each individual user's return signalan assigned-pilot may consume up to 20 percent of the total reversechannels air capacity.

Without phase and amplitude estimation, noncoherent or differentiallycoherent reception techniques must be performed. Accordingly, thereexists a need for a coherent demodulation system that reduces the aircapacity of the global-pilot and assigned-pilot signals whilemaintaining the desired air-interface performance.

SUMMARY

The present invention relates to a digital spread spectrum communicationsystem that employs pilot-aided coherent multipath demodulation with asubstantial reduction in global-pilot and assigned-pilot overheads. Thesystem and method uses a QPSK-modulated data signal whereby themodulated data is removed and the recovered carrier is used for channelamplitude and phase estimation. The resulting signal has no datamodulation and is used as a pseudo-pilot signal. In conjunction with thepseudo-pilot signal, a multiple-input phase-locked loop is employedfurther eliminating errors due to carrier-offset by using a plurality ofpseudo-pilot signals. A pilot signal is still required to resolveabsolute phase ambiguity, but at a greatly reduced magnitude.

Accordingly, it is an object of the present invention to provide a codedivision multiple access communication system which reduces the requiredglobal and assigned-pilot signal strength.

It is a further object of the invention to reduce the transmitted levelsof the global and assigned-pilots such that they consume negligibleoverhead in the air interface while providing information necessary forcoherent demodulation.

Other objects and advantages of the system and method will becomeapparent to those skilled in the art after reading the detaileddescription of the preferred embodiment.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified block diagram of a typical, prior art, CDMAcommunication system.

FIG. 2 is a detailed block diagram of a B-CDMA™ communication system.

FIG. 3A is a plot of an in-phase bit stream.

FIG. 3B is a plot of a quadrature bit stream.

FIG. 3C is a plot of a pseudo-noise (pn) bit sequence.

FIG. 4 is a detailed block diagram of the present invention using onepseudo-pilot signal, with carrier-offset correction implemented at thechip level.

FIG. 5 is a block diagram of a rake receiver.

FIG. 6 is a diagram of a received symbol p_(o) on the QPSK constellationshowing a hard decision.

FIG. 7 is a diagram of the angle of correction corresponding to theassigned symbol.

FIG. 8 is a diagram of the resultant symbol error after applying thecorrection corresponding to the assigned symbol.

FIG. 9 is a block diagram of a conventional phase-locked loop.

FIG. 10 is a detailed block diagram of the present invention using apseudo-pilot signal with carrier-offset correction implemented at thesymbol level.

FIG. 11 is a detailed block diagram of the present invention using apseudo-pilot signal and the MIPLL, with carrier-offset correctionimplemented at the chip level.

FIG. 12 is a block diagram of the multiple input phase-locked loop(MIPLL).

FIG. 13 is a detailed block diagram of the present invention using apseudo-pilot signal and the MIPLL, with carrier-offset correctionimplemented at the symbol level.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The preferred embodiment will be described with reference to the drawingfigures where like numerals represent like elements throughout.

A B-CDMA™ communication system 25 as shown in FIG. 2 includes atransmitter 27 and a receiver 29, which may reside in either a basestation or a mobile user receiver. The transmitter 27 includes a signalprocessor 31 which encodes voice and nonvoice signals 33 into data atvarious rates, e.g. data rates of 8 kbps, 16 kbps, 32 kbps, or 64 kbps.The signal processor 31 selects a rate in dependence upon the type ofsignal, or in response to a set data rate.

By way of background, two steps are involved in the generation of atransmitted signal in a multiple access environment. First, the inputdata 33 which can be considered a bi-phase modulated signal is encodedusing forward error-correcting coding (FEC) 35. For example, if a R=½convolution code is used, the single bi-phase modulated data signalbecomes bivariate or two bi-phase modulated signals. One signal isdesignated the in-phase channel I 41 a. The other signal is designatedthe quadrature channel Q 41 b. A complex number is in the form a+bj,where a and b are real numbers and j²=−1. Bi-phase modulated I and Qsignals are usually referred to as quadrature phase shift keying (QPSK).In the preferred embodiment, the tap generator polynomials for aconstraint length of K=7 and a convolutional code rate of R=½ areG₁=171₈ 37 and G₂=133₈ 39.

In the second step, the two bi-phase modulated data or symbols 41 a, 41b are spread with a complex pseudo-noise (pn) sequence. The resulting I45 a and Q 45 b spread signals are combined 53 with other spread signals(channels) having different spreading codes, multiplied (mixed) with acarrier signal 51, and transmitted 55. The transmission 55 may contain aplurality of individual channels having different data rates.

The receiver 29 includes a demodulator 57 a, 57 b which mixes down thetransmitted broadband signal 55 into an intermediate carrier frequency59 a, 59 b. A second down conversion reduces the signal to baseband. TheQPSK signal is then filtered 61 and mixed 63 a, 63 b with the locallygenerated complex pn sequence 43 a, 43 b which matches the conjugate ofthe transmitted complex code. Only the original waveforms which werespread by the same code at the transmitter 27 will be effectivelydespread. Others will appear as noise to the receiver 29. The data 65 a,65 b is then passed onto a signal processor 59 where FEC decoding isperformed on the convolutionally encoded data.

As shown in FIGS. 3A and 3B, a QPSK symbol consists of one bit each fromboth the in-phase (I) and quadrature (Q) signals. The bits may representa quantized version of an analog sample or digital data. It can be seenthat symbol duration t_(s) is equal to bit duration.

The transmitted symbols are spread by multiplying the QPSK symbol streamby a unique complex pn sequence. Both the I and Q pn sequences arecomprised of a bit stream generated at a much higher rate, typically 100to 200 times the symbol rate. One such pn sequence is shown in FIG. 3C.The complex pn sequence is mixed with the complex-symbol bit streamproducing the digital spread signal. The components of the spread signalare known as chips having a much smaller duration t_(c).

When the signal is received and demodulated, the baseband signal is atthe chip level. Both the I and Q components of the signal are despreadusing the conjugate of the pn sequence used during spreading, returningthe signal to the symbol level. However, due to carrier-offset, phasecorruption experienced during transmission manifests itself bydistorting the individual chip waveforms. If carrier-offset correctionis performed at the chip level, it can be seen that overall accuracyincreases due to the inherent resolution of the chip-level signal.Carrier-offset correction may also be performed at the symbol level, butwith less overall accuracy. However, since the symbol rate is much lessthan the chip rate, less overall processing speed is required when thecorrection is done at the symbol level.

System architectures for receivers taught in accordance with the systemand method of the present invention that do not require large magnitudepilot signals follow. The following systems replace the filtering,despreading and signal processing shown in FIG. 2. The systems areimplemented with carrier-offset correction at both the chip and symbollevels.

As shown in FIG. 4, a receiver using the system 75 and method of thepresent invention is shown. A complex baseband digital spread spectrumsignal 77 comprised of in-phase and quadrature phase components is inputand filtered using an adaptive matched filter (AMF) 79 or other adaptivefiltering means. The AMF 79 is a transversal filter (finite impulseresponse) which uses filter coefficients 81 to overlay delayed replicasof the received signal 77 onto each other to provide a filtered signal83 having an increased signal-to-noise ratio (SNR). The output 83 of theAMF 79 is coupled to a plurality of channel despreaders 85 ₁, 85 ₂, 85_(n) and a pilot despreader 87. In the preferred embodiment, n=3. Thepilot signal 89 is despread with a separate despreader 87 and pnsequence 91 contemporaneous with the transmitted data 77 assigned tochannels which are despread 85 ₁, 85 ₂, 85 _(n) with pn sequences 93 ₁,93 ₂, 93 _(n) of their own. After the data channels are despread 85 ₁,85 ₂, 85 _(n), the data bit streams 95 ₁, 95 ₂, 95 _(n) are coupled toViterbi decoders 97 ₁, 97 ₂, 97 _(n) and output 99 ₁, 99 ₂, 99 _(n).

The filter coefficients 81, or weights, used in adjusting the AMF 79 areobtained by the demodulation of the individual multipath propagationpaths. This operation is performed by a rake receiver 101. The use of arake receiver 101 to compensate for multipath distortion is well knownto those skilled in the communication arts.

As shown in FIG. 5, the rake receiver 101 consists of a parallelcombination of path demodulators (“fingers”) 103 ₀, 103 ₁, 103 ₂, 103_(n) which demodulate a particular multipath component. The pilotsequence tracking loop of a particular demodulator is initiated by thetiming estimation of a given path as determined by a pn sequence 105. Inthe prior art, a pilot signal is used for despreading the individualsignals of the rake. In this embodiment of the present invention, the pnsequence 105 may belong to any channel 93 ₁ of the communication system.The channel with the largest received signal is typically used.

Each path demodulator includes a complex mixer 107 ₀, 107 ₁, 107 ₂, 107_(n), and summer and latch 109 ₀, 109 ₁, 109 ₂, 109 _(n). For each rakeelement, the pn sequence 105 is delayed τ 111 ₁, 111 ₂, 111 _(n) by onechip and mixed 107 ₁, 107 ₂, 107 _(n) with the baseband spread spectrumsignal 113 thereby despreading each signal. Each multiplication productis input into an accumulator 109 ₀, 109 ₁, 109 ₂, 109 _(n) where it isadded to the previous product and latched out after the nextsymbol-clock cycle. The rake receiver 101 provides relative path valuesfor each multipath component. The plurality of n-dimension outputs 115₀, 115 ₁, 115 ₂, 115 _(n) provide estimates of the sampled channelimpulse response that contain a relative phase error of either 0°, 90°,180°, or 270°.

Referring back to FIG. 4, the plurality of outputs from the rakereceiver are coupled to an n-dimensional complex mixer 117. Mixed witheach rake receiver 101 output 115 is a correction to remove the relativephase error contained in the rake output.

A pilot signal is also a complex QPSK signal, but with the quadraturecomponent set at zero. The error correction 119 signal of the presentinvention is derived from the despread channel 95 ₁ by first performinga hard decision 121 on each of the symbols of the despread signal 95 ₁.A hard decision processor 121 determines the QPSK constellation positionthat is closest to the despread symbol value.

As shown in FIG. 6, the Euclidean distance processor compares a receivedsymbol p_(o) of channel 1 to the four QPSK constellation points x_(1,1),x_(−1,1), x_(−1,−1) , x_(1,−1). It is necessary to examine each receivedsymbol p_(o) due to corruption during transmission 55 by noise anddistortion, whether multipath or radio frequency. The hard decisionprocessor 121 computes the four distances d₁, d₂, d₃, d₄ to eachquadrant from the received symbol p_(o) and chooses the shortestdistance d₂ and assigns that symbol location x_(−1,1). The originalsymbol coordinates p_(o) are discarded.

Referring back to FIG. 4, after undergoing each hard symbol decision121, the complex conjugates 123 for each symbol output 125 aredetermined. A complex conjugate is one of a pair of complex numbers withidentical real parts and with imaginary parts differing only in sign.

As shown in FIG. 7, a symbol is demodulated or derotated by firstdetermining the complex conjugate of the assigned symbol coordinatesx_(−1,−1), forming the correction signal 119 which is used to remove therelative phase error contained in the rake output. Thus, the rake outputis effectively derotated by the angle associated with the hard decision,removing the relative phase error. This operation effectively provides arake that is driven by a pilot signal, but without an absolute phasereference.

Referring back to FIG. 4, the output 119 from the complex conjugate 123is coupled to a complex n-dimensional mixer 117 where each output of therake receiver 101 is mixed with the correction signal 119. The resultingproducts 127 are noisy estimates of the channel impulse response p₁ asshown in FIG. 8. The error shown in FIG. 8 is indicated by a radiandistance of π/6 from the in-phase axis.

Referring back to FIG. 4, the outputs 129 of the complex n-dimensionalmixer 117 are coupled to an n-dimensional channel estimator 131. Thechannel estimator 131 is a plurality of low-pass filters filtering eachmultipath component. The outputs of the n-dimensional mixer 117 arecoupled to the AMF 79. These signals act as the AMF 79 filter weights.The AMF 79 filters the baseband signal to compensate for channeldistortion due to multipath without requiring a large magnitude pilotsignal.

Rake receivers 101 are used in conjunction with phase-locked loop (PLL)133 circuits to remove carrier-offset. Carrier-offset occurs as a resultof transmitter/receiver component mismatches and other RF distortion.The present invention 75 requires that a low level pilot signal 135 beproduced by despreading 87 the pilot from the baseband signal 77 with apilot pn sequence 91. The pilot signal is coupled to a single input PLL133. The PLL 133 measures the phase difference between the pilot signal135 and a reference phase of 0. The despread pilot signal 135 is theactual error signal coupled to the PLL 133.

A conventional PLL 133 is shown in FIG. 9. The PLL 133 includes anarctangent analyzer 136, complex filter 137, an integrator 139 and aphase-to-complex-number converter 141. The pilot signal 135 is the errorsignal input to the PLL 133 and is coupled to the complex filter 137.The complex filter 137 includes two gain stages, an integrator 145 and asummer 147. The output from the complex filter is coupled to theintegrator 139. The integral of frequency is phase, which is output 140to the converter 141. The phase output 140 is coupled to a converter 141which converts the phase signal into a complex signal for mixing 151with the baseband signal 77. Since the upstream operations arecommutative, the output 149 of the PLL 133 is also the feedback loopinto the system 75.

By implementing the hard decision 121 and derotation 123 of the datamodulation, the process provides channel estimation without the use of alarge pilot signal. If an error occurs during the hard decision processand the quadrant of the received data symbol is not assigned correctly,the process suffers a phase error. The probability of phase error isreduced, however, due to the increased signal-to-noise ratio of thetraffic channel. The errors that occur are filtered out during thechannel-estimation and carrier-recovery processes. The traffic channelis approximately 6 dB stronger (2×) than the level of the despreadpilot.

As described earlier, the present invention can also be performed withcarrier-offset correction at the symbol level. An alternative embodiment150 implemented at the symbol level is shown in FIG. 10. The differencebetween the chip and symbol level processes occur where the output ofthe conventional PLL 133 is combined. At the symbol level, the PLLoutput 140 does not undergo chip conversion 141 and is introduced intothe AMF 79 weights after the rake receiver 101 by another n-dimensionalmixer 153. The phase correction 140 feedback must also be mixed 154 ₁,154 ₂, 154 _(n) with the outputs 95 ₁, 95 ₂, 95 _(n) of each of theplurality of channel despreaders 85 ₁, 85 ₂, 85 _(n) and mixed 156 withthe output 135 of the pilot despreader 87.

As shown in FIG. 11, another alternative embodiment 193 uses a variationof the earlier embodiments whereby a hard decision is rendered on eachreceived symbol after despreading and derotated by a radian amount equalto the complex conjugate. The alternate approach 193 uses a plurality ofchannel despreaders 85 ₁, 85 ₂, 85 _(n) and the pilot despreader 87 asinputs to a multiple input phase-locked loop (MIPLL) 157 shown in FIG.12. Since each of the despread channels 95 ₁, 95 ₂, 95 _(n) contains anambiguous representation of the pilot signal, a small signal pilot 135is required to serve as an absolute reference. The despread symbols fromall channels in conjunction with the despread small signal pilot signalare input to the MIPLL 157.

Referring to FIG. 12, the output from each channel 95 ₁, 95 ₂, 95 _(n)is coupled to a hard decision/complex conjugate operation 159 ₁, 159 ₂,159 _(n). The derotated pseudo-pilots 161 ₁, 161 ₂, 161 _(n) are thenmixed with the delayed symbols producing a complex voltage error 163 ₁,163 ₂, 163 _(n). The error 165 ₁, 165 ₂, 165 _(n) is input into aconverter 167 ₁, 167 ₂, 167 _(n), 167 _(n+1), which takes an inversetangent converting the complex number into a phase error 169 ₁, 169 ₂,169 _(n), 169 _(n+1). Each phase error 169 ₁, 169 ₂, 169 _(n), 169_(n+1) is input into a maximum likelihood combiner 171 which assignsvarious weights to the plurality of inputs and produces a sum output.Also included in the sum is the small signal pilot 135 phase 169 _(n+1)which is despread 135 and converted 167 _(n+1). The weighting of thesmall pilot signal may be emphasized since its phase is unambiguous.

The output of the combiner 173 is the estimate of the carrier-offset andis coupled to a complex filter 175 and coupled to an integrator 177. Allchannels contribute to the estimate of the carrier-offset frequency withthe absolute phase error removed by the unambiguous pilot signal. Theintegrator accumulates the history of the summed signal over manysamples. After integration, the estimate of the phase error is output179 converted to a complex voltage and output 183.

Referring back to FIG. 11, the output 183 of the MIPLL 157 is coupled toa complex mixer 185 upstream of the rake receiver. This completes theerror feedback for the MIPLL 157. Even though this embodiment requiresadditional resources and complexity, the MIPLL 157 architecture can beefficiently implemented and executed in a digital signal processor(DSP).

Referring now to the alternative embodiment 195 shown in FIG. 13, thisembodiment 195 mixes the output of the MIPLL 157 at the symbol level.The MIPLL 157 is mixed 197 with the output of the rake receiver 101. Asdescribed above, the output of the rake receiver 101 is at the symbollevel. The symbol-to-chip conversion 181 in the MIPLL 157 architectureis disabled. Since the output 183 of the MIPLL 157 is mixed with theoutputs of the rake 101 which are used only for the AMF 79 weights, thephase correction for carrier-offset must be added to the portion of thereceiver that processes traffic data. A plurality of mixers 199 ₁, 199₂, 199 _(n) downstream of each channel despreader 85 ₁, 85 ₂, 85 _(n)and a mixer 193 downstream of the pilot despreader 87 are thereforerequired to mix the phase-corrected output 183 (at the symbol level) asfeedback into the system.

The present invention maintains the transmitted pilot signal at a lowlevel to provide an absolute phase reference while reducing pilotinterference and increasing air capacity. The net effect is the virtualelimination of the pilot overhead.

While specific embodiments of the present invention have been shown anddescribed, many modifications and variations could be made by oneskilled in the art without departing from the spirit and scope of theinvention. The above description serves to illustrate and not limit theparticular form in any way.

What is claimed is:
 1. A receiver for use in a communication station ofa CDMA system wherein a plurality of communication stations communicatewith each other over a CDMA air interface using a plurality of channelsand a pilot signal for carrier-offset recovery during reception, thereceiver comprising: an adaptive matched filter for receivingdemodulated CDMA communication signals producing a filtered signal byusing a weighting signal; a rake receiver for receiving the demodulatedCDMA communication signals and a pseudo-noise signal generated for aselected channel and producing a filter weighting signal; means fordefining the filter weighting signal with a correction signal, saidcorrection signal to produce the weighting signal used by said adaptivematched filter; a channel despreader for said selected channel coupledto said adaptive matched filter output for despreading said filteredsignal using the pseudo-noise signal generated for said selected channelto produce a despread channel signal of said selected channel; a pilotchannel despreader for a pilot channel coupled to said adaptive matchedfilter output for despreading said filtered signal using a pseudo-noisesignal generator for said pilot channel to produce a despread pilotsignal of said pilot channel; a hard decision processor in associationwith a complex conjugate processor for receiving the despread channelsignal of said selected channel and producing said correction signal;and a phase-locked loop utilizing at least said despread pilot signalfor producing a phase correction signal which is applied to producephase-corrected channel signals.
 2. The receiver according to claim 1further comprising a plurality of channel despreaders, each coupled tosaid adaptive matched filter output for despreading said filtered signaleach using an associated pseudo-noise signal generator to produce aplurality of despread channel signals.
 3. The receiver according toclaim 2 wherein the number of channel despreaders is three.
 4. Thereceiver according to claim 2 wherein said phase-locked loop phasecorrection signal is at a chip level and is applied to said demodulatedCDMA communication signals.
 5. The receiver according to claim 2 whereineach of the plurality of channels is a complex, bi-phase modulatedsignal comprised of symbols including in-phase and quadrature componentsrepresenting data, said hard decision processor compares each despreadchannel signal symbol to one of four possible quadrature constellationpoints and assigns each of said symbols to a nearest constellationpoint, and said complex conjugate processor derotates each of saidsymbols by determining a complex conjugate of each of said assignedpoints to produce said correction signal.
 6. The receiver according toclaim 2 wherein said phase-locked loop further comprises a plurality ofinputs corresponding with said plurality of channel despreaders.
 7. Thereceiver according to claim 6 wherein said phase-locked loop furthercomprises: a hard decision processor in association with said complexconjugate processor with a local feedback loop for each of saidcorresponding channel despreader inputs to produce an error estimatesignal for a respective channel signal; each said error estimate signaland said despreader pilot signal coupled to an inverse tangent processorto produce a corresponding phase correction signal; and said respectivechannel phase correction signal and pilot phase correction signalcoupled to a maximum likelihood combiner producing a combinationcorrection signal coupled to an integrator to produce said phasecorrection signal.
 8. The receiver according to claim 7 wherein thenumber of channel despreaders is three.
 9. The receiver according toclaim 1 wherein said phase-locked loop phase correction signal is at asymbol level and is applied to said filter weighting signal and to saiddespread channel signals of said channel and pilot channel despreaders.10. The receiver according to claim 9 further comprising a plurality ofchannel despreaders, each coupled to said adaptive matched filter outputfor despreading said filtered signal using an associated pseudo-noisesignal generator to produce a plurality of despread channel signals. 11.The receiver according to claim 10 wherein the number of channeldespreaders is three.
 12. The receiver according to claim 10 whereinsaid phase-locked loop further comprises a plurality of signal inputscorresponding with said plurality of channel despreaders.
 13. Thereceiver according to claim 12 wherein said phase-locked loop furthercomprises: a hard decision processor in association with a complexconjugate processor with a local feedback loop for each of saidplurality of signal inputs, each producing an error estimate for arespective channel signal; each of said channel error estimates and saiddespreader pilot signal coupled to an inverse tangent processoroutputting a channel phase correction signal; and said channel phasecorrection signal and said pilot phase correction signal coupled to amaximum likelihood combiner producing a combination correction signalcoupled to an integrator to produce said phase correction signal. 14.The receiver according to claim 13 wherein the number of channeldespreaders is three.
 15. A method of receiving at least one of aplurality of channels over a CDMA air interface using a reducedmagnitude pilot signal for carrier-offset recovery during receptionwherein a plurality of communication stations communicate with eachother comprising the steps: receiving demodulated CDMA communicationsignals; filtering said received demodulated CDMA communication signalswith an adaptive matched filter to produce a filtered signal by using aweighting signal; producing a filter weighting signal with a rakereceiver using said demodulated CDMA communication signals and apseudo-noise signal generated for a selected channel; refining saidfilter weighting signal with a correction signal; despreading saidselected channel from said filtered signal using the pseudo-noise signalfor said selected channel to produce a despread channel signal of saidselected channel; despreading a pilot channel from said filtered signalusing a pseudo-noise signal generated for said pilot channel to producea despread pilot signal of said pilot channel; processing said selecteddespread channel signal with a hard decision processor in associationwith a complex conjugate processor to produce said correction signal;and generating a phase correction signal from said despread pilot signalwith a phased-locked loop to phase-correct said selected channel signal.16. The method according to claim 15 wherein said phase correctionsignal is at a chip level.
 17. The method according to claim 16 whereinthe step of despreading said selected channel also includes despreadinga plurality of channels to produce despread channel signals.
 18. Themethod according to claim 17 wherein said step of generating a phasecorrection signal further includes the steps of: assigning a receivedsymbol to one of four possible quadrature constellation points for saiddespread selected channel signal and each of said despread channelsignals; derotating each of said assigned symbols for said despreadselected channel signal and each of said despread channel signals bydetermining the complex conjugate of each of said assigned points toproduce respective error estimate signals; coupling each of said errorestimate signals and said despread pilot signal to inverse tangentprocessors to produce corresponding phase correction signals; andcombining said channel phase correction signal and said pilot phasecorrection signal to produce said phase correction signal.
 19. Themethod according to claim 15 wherein said phase correction signal is ata symbol level.
 20. The method according to claim 19 wherein the step ofdespreading said selected channel also includes despreading a pluralityof channels to produce despread channel signals.
 21. The methodaccording to claim 20 wherein said step of generating a phase correctionsignal further includes the steps of: assigning a received symbol to oneof four possible quadrature constellation points for said despreadselected channel signal and each of said despread channel signals;derotating each of said assigned symbols for said despread selectedchannel signal and each of said despread channel signals by determiningthe complex conjugate of each of said assigned points to producerespective error estimate signals; coupling of each said error estimatesignals and said despread pilot signal to inverse tangent processors toproduce corresponding phase correction signals; and combining saidchannel phase correction signal and said pilot phase correction signalto produce said phase correction signal.